Suspended transmission line with embedded signal channeling device

ABSTRACT

A suspended transmission line with an embedded signal channeling device includes a support layer and a conductor supported by the support layer between first and second plates each having a ground plane. The conductor includes a combined signal line and a plurality of discrete signal lines extending from the combined signal line. The discrete signal lines each transmit a portion of a signal transmitted on the combined signal line. A propagation structure is disposed between the first and second plates to substantially contain an electromagnetic field generated by the propagating signal.

RELATED APPLICATIONS

This application is related to U.S. patent application Ser. No.09/548,691 entitled “Suspended Transmission Line and Method”, U.S.patent application Ser. No. 09/548,467 entitled “Suspended TransmissionLine with Embedded Amplifier”, U.S. application Ser. No. 09/548,578entitled “Integrated Broadside Conductor for Suspended Transmission Lineand Method”, and U.S. application Ser. No. 09/548,689 entitled “Methodfor Fabricating Suspended Transmission Line”, all filed on Apr. 13,2000.

TECHNICAL FIELD OF THE INVENTION

This invention relates generally to the field of signal transmissionsystems, and more particularly to a suspended transmission line with anembedded signal channeling device.

BACKGROUND OF THE INVENTION

Microwave and radio frequency circuits are generally implemented byinterconnecting amplifiers, antennas, transmitters, receivers, and othercomponents by a series of transmission lines. The transmission linespropagate microwave and radio frequency energy between the components ofthe circuit.

Transmission lines are generally implemented as waveguide pipes,striplines, and/or coaxial cables. Waveguide pipes are oftenimpractical, however, because of the difficulty of installation and thesize and weight is excessive for many applications. Striplines andcoaxial cables are more compact and easier to install, but use specialmaterials and fabrication processes that lead to high transmission linecost.

Further adding to the expense of microwave and radio frequency circuitsis the expense of implementing amplifiers, antennas, splitters,combiners, and other components within the circuit. Typically, eachcomponent is implemented in a specially fabricated mechanical housingsuch as an aluminum box having signal, digital, and power connectors.These mechanical housings must generally be designed, engineered, andmachined with tight tolerances for microwave and other high frequencyapplications. In addition, drawing packages need to be generated andmaintained for each application. Connectors must also be thermallymatched to the mechanical housing.

SUMMARY OF THE INVENTION

The present invention provides a transmission line signal channelingdevice that substantially eliminates or reduces the problems anddisadvantages associated with prior methods and systems. In particular,the signal channeling device is embedded into a suspended transmissionline to divide or combine signals in cellular and other suitablefrequency applications.

In accordance with one embodiment of the present invention, a suspendedtransmission line with an embedded signal channeling device includes asupport layer and a conductor supported by the support layer betweenfirst and second plates each having a ground plane. The conductorincludes a combined signal line and a plurality of discrete signal linesextending from the combined signal line. The discrete signal lines eachtransmit a portion of a signal transmitted on the combined signal line.A propagation structure is positioned between the first and secondplates to substantially contain an electromagnetic field generated bythe propagating signal.

More specifically, in accordance with a particular embodiment of thepresent invention, the discrete signal lines include first and secondoutside lines and a center line between the first and second outsidelines. The center line includes a serpentine element to maintainsubstantially the same length, and thus phase, as the first and secondoutside lines.

Technical advantages of the present invention include providing a lowcost and space efficient transmission system. In particular, a signalchanneling device is embedded into a transmission line substantiallywithout degradation in the height of the transverse cross-section of theline. As a result, the transmission line structure is self-contained andincludes fewer parts. The cost of designing, engineering, constructing,and maintaining separate mechanical housings for a divider or a combineris minimized.

In accordance with the present invention there is provided an improvedsignal channeling device for cellular and other suitable frequencyapplications. In particular, the signal channeling device is embeddedwithin and integral with a suspended transmission line. The resultingchanneling device provides good isolation and VSWR, as well as lowinsertion loss. In addition, the number of transmission line connectorsfor the channeling device is reduced. Thus, antenna and other systemsusing dividers and/or combiners may be constructed at relatively lowcost.

Other technical advantages of the present invention will be readilyapparent to one skilled in the art from the following figures,description, and claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention and itsadvantages, reference is now made to the following description taken inconjunction with the accompanying drawings, wherein like referencenumerals represent like parts, in which:

FIG. 1 is a sectional diagram illustrating a suspended transmission linein accordance with one embodiment of the present invention;

FIG. 2 is a sectional diagram illustrating distribution of an electricfield in the suspended transmission line of FIG. 1;

FIG. 3 is a flow diagram illustrating a method for fabricating thesuspended transmission line of FIG. 1 in accordance with one embodimentof the present invention;

FIG. 4 is a flow diagram illustrating a method for transmitting a signalin the transmission line of FIG. 1 in accordance with one embodiment ofthe present invention;

FIG. 5 is a perspective diagram illustrating a signal channeling deviceembedded into a suspended transmission line segment in accordance withone embodiment of the present invention;

FIG. 6 is a schematic diagram illustrating the configuration of thecenter conductor for the embedded signal channeling device of FIG. 5 inaccordance with one embodiment of the present invention;

FIG. 7 is a schematic diagram illustrating further details of thesuspended transmission line in accordance with one embodiment of thepresent invention; and

FIG. 8 is a flow diagram illustrating a method for dividing a signal ina transmission line in accordance with one embodiment of the presentinvention.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 illustrates a suspended transmission line 10 in accordance withone embodiment of the present invention. In this embodiment, thesuspended transmission line 10 is used to transmit microwave and otherradio frequency signals in a transmission system. As described in moredetail below, an amplifier, power divider, or other active or passivedevice may be embedded into the transmission line 10 to manipulate atransmitted signal. The transmitted signal may be an outgoing signalbeing transmitted to an antenna or incoming signal being received froman antenna. It will be understood that the suspended transmission line10 may be otherwise suitably configured for use in microwave, radiofrequency and other suitable high power or other applications.

Referring to FIG. 1, the suspended transmission line 10 includes asupport layer 12 supporting a center conductor 14, first and secondspacers 16 and 18 each disposed on opposite sides of the support layer12, and first and second plates 20 and 22 each disposed outwardly of acorresponding spacer 16 or 18. As described in more detail below, eachof the layers 12, 16, 18, 20, and 22 may be separately fabricated andthereafter laminated together to form the suspended transmission line10.

The support layer 12 is a thin dielectric sheet having a first side 24and an opposite second side 26. The thickness of the support layer 12 ispreferably minimized to a thickness needed to support the centerconductor 14 in order to minimize the cross section of the support layer12 and thus limit electrical fields in the layer 12. The support layer12 may be continuous or include openings (not shown) to controlpropagation characteristics of the suspended transmission line 10.Layers 20 and 22 may also contain holes to allow integration ofcomponents directly into the suspended transmission line 10, and thelike.

The support layer 12 is fabricated from an inexpensive dielectricmaterial which may have a moderate loss tangent as in a lossy material.The use of a lossy dielectric material for the support layer 12 avoidsthe necessity of exotic low-loss materials such as Alumina, Duroid,cross-linked polystyrene, and Beryllium Oxide previously used to supporta conductor in a suspended transmission line. Although such low-lossmaterials improve insertion loss, such materials are typically veryexpensive. A lossy material is the material of preference because, as adescribed in more detail below, the center conductor 14 is configured todirect an electric field generated by a signal on the center conductor14 substantially away from the support layer 12 such that only fringingelectrical fields cross the support layer 12. Dissipation losses due tothe fringing electrical fields are minimal even in the lossy material ofthe support layer 12. As a result, the suspended transmission line 10may be produced at relatively low cost and used in high power and highperformance applications.

The lossy material of the support layer 12 is an epoxy glass such asG-10 or GFG, polyimide glass, or other suitable printed circuit boardbase materials such as polyester, or other suitable lossy materials. Alossy material has a moderate loss tangent of about 0.04 or less. In oneembodiment, G-10 material is preferred for the support layer 12 becauseG-10 has good dimensional stability over a large temperature range andis easy to laminate and match to other layers and materials. In anotherembodiment, an incremental increase in performance is obtained by usinglow loss PTFE material in place of the G-10 for the support layer 12.Because the support layer 12 is thin, this results in only a smallincrease in cost.

The center conductor 14 is supported by the support layer 12 between thefirst and second plates 20 and 22. The first and second plates 20 and 22provide the upper and lower plates and act as ground planes to thesuspended transmission line 10. Plates 20 and 22 may be solid metal or abase substrate material with metal layers on both sides. The centerconductor 14 transmits the signal with low dissipation loss.Accordingly, the suspended transmission line 10 has utility to carry asignal over long distances between amplifiers, antennas, transmitters,receivers, and other components in the transmission or receiver system.

The center conductor 14 includes a first part 34 exposed at the firstside 24 of the support layer and a second part 36 exposed at the secondside 26 of the support layer 12. The first and second parts 34 and 36 ofthe center conductor 14 preferably mirror each other to minimize in thesupport layer 12 the electric field generated by a signal transmitted onthe center conductor 14.

A third part 38 of the center conductor 14 connects the first and secondparts 34 and 36 at intermediate points 40 along the length of the centerconductor 14. Connection of the first and second parts 34 and 36 at theintermediate points 40 produces equal phase and amplitude for a signalbetween the first and second parts 34 and 36 and reduces electric fieldcoupling. As a result of this structure, the electric field generated bya transmitted signal is substantially directed away from the supportlayer 12 with only fringing electric fields in the support layer 12.Further details of a typical electric field distribution are describedbelow in connection with FIG. 2.

Connection of the first and second parts 34 and 36 of the centerconductor 14 at the intermediate points 40 means the first and secondparts 34 and 36 are electrically connected to each other at least atspaced intervals along the length of the center conductor 14. Spacingbetween the intermediate points 40 is substantially equal along thelength of the center conductor 14 and is based on the frequency of thesignal to be transmitted by the suspended transmission line 10. In aparticular embodiment, the center conductor 14 includes about 10 to 20connections per wavelength of the transmitted signal frequency. It willbe understood that other suitable spacing that maintains a substantiallyconstant phase and amplitude for a signal on the center conductor 14 maybe used.

In the illustrated embodiment, the center conductor 14 is an integratedbroadside conductor. For this embodiment, the first part 34 of thecenter conductor 14 is a first conductive strip 44 disposed on the firstside 24 of the support layer and the second part 36 of the centerconductor 14 is a second conductive strip 46 disposed on the second side26 of the support layer 12. The first and second conductive strips 44and 46 are copper or silver-plated copper or other suitable metal tracesthat minimize conductor resistivity. The third part 38 of the centerconductor 14 comprises a plurality of broadside connectors 48 eachextending through the support layer 12 between the first and secondconductive strips 44 and 46 to electrically couple the strips 44 and 46at an intermediate point 40. Unless otherwise specified, the use of theterm each herein means each of at least a subset of the identifieditems. The connectors 48 are copper or silver-plated copper vias orother suitable conductive connectors.

The first and second spacers 16 and 18 maintain the plates 20 and 22 inspace relation with the support layer 12, and thus the center conductor14, to form a propagation structure 50 encompassing the center conductor14 with air and ground planes for Quasi-TEM mode of propagation. Thepropagation structure 50 encompasses the center conductor 14 in that itis over, including above and/or below the conductor 14 up to and beyondthe upper and lower ground plates 20 and 22. As described in more detailbelow, the propagation structure 50 provides a low-loss medium forpropagation of the electromagnetic field generated by a transmittedsignal. Accordingly, dissipation losses are minimized along thesuspended transmission line 10.

The first and second spacers 16 and 18 may each be continuous along thepropagation structure 50 or comprise a plurality of discrete posts orother suitable structures operable to maintain the plates 20 and 22 inspace relation from the center conductor 14. The spacers 16 and 18 aresized such that substantially all of the electromagnetic field generatedby a transmitted signal around the center conductor 14 is maintained inthe propagation structure 50. Thus, as described in more detail below,spacer geometry is dependent on the transmitted signal frequency as wellas the size, geometry, and materials of the support layer 12, centerconductor 14, plates 20 and 22, and propagation structure 50.

The first and second spacers 16 and 18 are each fabricated of adielectric, conductor, or other suitable material or materials.Preferably, the sidewalls of the spacers 16 and 18 are spaced apart andaway from the center conductor 14 to minimize the effect on theelectromagnetic field in the propagation structure 50. This minimizesthe changes in impedance along the direction of propagation. Inaddition, the spacer material preferably has a coefficient of thermalexpansion equal or at least similar to the material of the support layer12 so that the suspended transmission line 10 has good mechanicalstability over a large temperature range. In a particular embodiment,the support layer 12 and spacers 16 and 18 are each fabricated of G-10material.

For the illustrated embodiment, each spacer 16 and 18 includes adhesionlayers 60 at each edge for connecting the spacers to the support layer12 and the plates 20 and 22. The support layer 12 includes ametalization layer 62 on each side 24 and 26. The metalization layers 62form the point of attachment of the mode suppression connection vias 68.These minimize the impedance altering effects of the potential higherorder modes and reduce electromagnetic coupling between alternate boardroutes. In this embodiment, the first spacer 16 is attached to themetalization layer 62 on the first side 24 of the support layer 12 toseparate the first plate 20 from the center conductor 14. The secondspacer 18 is attached to the metalization layer 62 on the second side 26of the support layer 12 to separate the second plate 22 from the centerconductor 14. Both layers 62 on 24 and 26 are attached to the connectionvias 68.

The first and second plates 20 and 22 may consist entirely of conductivemetal such as copper or may consist of a lossy dielectric with copper orsilver-plated copper or other suitable metal with low resistivity oneach side. These plates form the ground plane 66 disposed over thecenter conductor 14. The ground planes 66 of the plates 20 and 22 andthe underlying conductive strips 44 and 46 of the center conductor 14together generate the electromagnetic field in the propagation structure50. Variations in spacing of the ground plane 66 from the centerconductor 14 may be offset by the line width 64 of the center conductor14 in order to maintain a substantially constant impedance in the centerconductor 14. Spacing variations may be caused by access openings 65 cutin the ground plane 66 to allow insertion and integration of a deviceinto the suspended transmission line 10. In this case, the accessopening is illustrated covered by a magnetic or other ground cover 67secured flush with the outside of the plate 20 or 22.

In the illustrated embodiment, the plates 20 and 22 comprise aconductive material and each form a continuous ground plane 66. In aparticular embodiment, the plates 20 and 22 are copper plates having athin outer tin layer (not shown) to reduce corrosion and improvesolderability. The first plate 20 is attached outwardly of the firstspacer 16 to form a first propagation cavity 70 between the first plate20 and the first conductive strip 44 of the center conductor 14. Thesecond plate 22 is attached outwardly of the second spacer 18 to form asecond propagation cavity 72 between the second plate 22 and the secondconductive strip 46 of the center conductor 14. In this embodiment, thefirst and second propagation cavities 70 and 72 form the propagationstructure 50.

The propagation cavities 70 and 72 each provide a low-loss medium forpropagation of the electromagnetic field generated by a transmittedsignal on the center conductor 14. The low-loss medium is a medium thatpropagates the electromagnetic field with a dissipation loss on theorder of about 0.1 dB/inch or below at microwave frequencies. In theillustrated embodiment, the propagation cavities 70 and 72 are each anair cavity. To prevent moisture from entering the suspended transmissionline 10, the propagation cavities 70 and 72 may include closed cell foamor other suitable low-loss material to displace the air and reduceoverall moisture content.

A plurality of mode suppression connectors 68 are formed on either sideof the propagation structure 50 to eliminate or reduce interferencebetween the suspended transmission line 10 and nearby or adjacenttransmission lines and other devices or circuits in the transmissionsystem. The mode suppression connectors 68 are spaced in accordance withconventional techniques. In one embodiment, the mode suppressionconnectors 68 are tin plated copper vias extending through the supportlayer 12 and spacers 16 and 18 between the plates 20 and 22. The modesuppression connectors 68 are attached to the metalization layers 62 foradditional mechanical support and improved mode suppression.

For the suspended transmission line 10, the geometry, size, and materialof the support, spacer, and plate layers 12, 16, 18, 20 and 22 and ofthe center conductor 14 and propagation cavities 70 and 72 are dependenton the frequency of a signal to be transmitted by the line 10. Therelationship between the maximum transmitted signal frequency and thematerials and geometry of the suspended transmission line 10 dictatethat the transmission line 10 should be operated below the first cut-offfrequency of the potential higher order modes. The onset of the firsthigher order mode may be approximated by frequency equations forrectangular waveguide and for non-integrated suspended stripline. Thefrequency equation for rectangular waveguide is as follows:$f_{c} = \frac{c0}{2 \cdot a}$

where:

f_(c)=TE10 mode cut-off frequency

c0=the speed of light in a vacuum

a=the enclosure (waveguide) width

The frequency equation for non-integrated suspended striplines is asfollows:$f_{c} = {\frac{c0}{2 \cdot a}{\sqrt{1 - \frac{h}{b}}\quad\left\lbrack \frac{e_{r} - 1}{e_{r}} \right\rbrack}}$

where:

f_(c)=1st higher order mode cut-off frequency

c0=the speed of light in vacuum

a=the enclosure width

b=the enclosure height

h=the supporting substrate height

e_(r)=the relative dielectric constant of the dielectric support layer

These equations provide a close estimation of values for the suspendedtransmission line 10, with the non-integrated suspended striplineequation generally providing a closer approximation due to the lowcut-off frequency of the suspended transmission line 10. Other factorshave been known to change the cut-off frequency of the suspendedtransmission line 10. Such factors include the size and spacing of themode suppression connectors 68 and the size and spacing of the broadsideconnectors 48.

In one embodiment, values obtained for a particular implementation ofthe suspended transmission line 10 from the equations may be fine tunedusing conventional computer simulation techniques and programs, modifiedto account for the configuration of the suspended transmission line 10.Suitable programs include ANSOFT EXTRACTOR for 2-D analysis and ANSOFTHFSS (High Frequency Structure Simulator) for 3-D analysis. Furthermodeling may be done using the BRCTL mode of the HP MDS LINECALC model.The LINECALC model is normally used for a pair of coupled lines asopposed to a single transmission line with mode voltages identical atany cross-section. In the BRCTL model, the even mode characteristicimpedance is used as twice the characteristic impedance of the suspendedtransmission line 10 to account for the dual parallel configuration ofthe center conductor 14. In addition, because the space between thefirst and second conductive strips 44 and 46 has nearly the samepotential as the first and second conductive strips 44 and 46 when theyare held at the same potential by the connectors 48, the first andsecond conductive strips 44 and 46 can be regarded as an approximationto a single thick conductor. Accordingly, the thickness of theconductive strips 44 and 46 must be accounted for in impedancecalculations. Other modes of the MDS LINECALC model, finite elementanalysis, and other suitable techniques are available to simulate anddesign the suspended transmission line 10.

In a particular microwave embodiment of the suspended transmission line10, the support layer 12 comprises G-10 material having a relativedielectric constant of 4.5 and a thickness of 8 mils. In this particularembodiment, the first and second spacers 16 and 18 each comprise G-10material and with the adhesion layers 60 having a thickness of 38 mils.The plates 20 and 22 are each copper and have a thickness of 20 mils.Each propagation cavity 70 and 72 has an enclosure width of 240 mils andan enclosure height of 38 mils. The center conductor 14 has an impedanceof 50 ohms and comprises of copper plated silver traces 44 and 46 oneach side 24 and 26 of the support layer and plated silver connectors 48extending through the support layer 12 between the silver-plated traces44 and 46. The silver-plated traces 44 and 46 each have a thickness of1.5 mils and a line width of 20 mils. The connectors 48 have a diameterof 13 mils and are each spaced approximately 100 mils apart. Thesuspended transmission line 10 of this embodiment has a cut offfrequency of 14.2 GHz and an upper useful range of about 14 GHz, whichprovides a margin to account for manufacturing tolerances. Testing ofthis suspended transmission line 10 showed an insertion loss of 0.02 dBper inch at 1 GHz, an insertion loss is 0.05 dB per inch at 5 GHz, aninsertion loss of 0.12 dB per inch at 10 GHz, and an insertion loss of0.55 dB per inch at 15 GHz.

FIG. 2 illustrates distribution of an electric field 80 in the suspendedtransmission line 10. As previously described, the electric field 80 isgenerated by a propagating electromagnetic signal around the centerconductor 14. This signal may be a microwave, radio, or other suitablefrequency signal.

Referring to FIG. 2, the electric field 80 includes a transverse field82 generally perpendicular to the conductive strips 44 and 46 and smallfringing fields 84 in the support layer 12 at the edges of theconductive strips 44 and 46. The transverse field 82 comprises thesignificant part of the electric field 80 and is propagated in thelow-loss medium of the propagation cavities 70 and 72. Accordingly,dissipation or insertion losses are minimal and limited to losses in thefringing field 84.

FIG. 3 is a flow diagram illustrating a method for fabricating thesuspended transmission line 10 of FIG. 1 in accordance with oneembodiment of the present invention. In this embodiment, the support,spacer, and ground layers 12, 16, 18, 20, and 22 are separatelyfabricated and thereafter laminated together to form the suspendedtransmission line 10. It will be understood that the suspendedtransmission line 10 may be otherwise fabricated and comprise othermaterials without departing from the scope of the present invention.

Referring to FIG. 3, the method begins at step 100 in which theconductive traces 44 and 46 are formed on each side 24 and 26 of thesupport layer 12. In one embodiment, copper on support layer 12 ispatterned and etched to form the conductive traces 44 and 46. The silveris plated on the resultant copper traces. In this embodiment, the silverplating may be applied outside or at the edges of the cavities 70 and 72to form the metalization layers 62 and provide an intermediate groundplane on each side 24 and 26 of the support layer 12. The support layer12 is cut to size before or after formation of the conductive traces 44and 46.

Proceeding to step 102, connectors 48 are formed in the support layer 12between the first and second conductive strips 44 and 46. In oneembodiment, the connectors 48 are formed by drilling vias at theintermediate points 40 along the conductive strips 44 and 46 and platingthe vias with copper. These vias are then silver plated. As previouslydescribed, the connectors 48 provide equal phase and amplitude for asignal between the first and second conductive strips 44 and 46 toreduce electric field coupling. The reduced electric field couplingbetween conductive strips 44 and 46 lead to reduced insertion loss.

Next, at step 104, the first and second spacers 16 and 18 are eachformed by a pair of opposing strips routed or otherwise formed from aspacer layer. The spacer layer preferably comprises an inexpensivematerial that is thermally matched to the support layer 12.

At step 106, the spacers 16 and 18 are each laminated to opposite sides24 and 26 of the support layer 12. The strips for each spacer arepositioned along edges of the support layer 12 and displaced from thecenter conductor 14 to form the propagation cavities 70 and 72. In oneembodiment, the spacers 16 and 18 are laminated to the support layer 12using a conventional no-flow or low-flow stage process. In the B-stageprocess, partially cured epoxy with glass cloth reinforcement isattached to the support layer 12 and becomes part of overlying spacer 16or 18. The geometry of the glass cloth reinforcement matches that of theoverlying spacer 16 or 18 so as to not interfere with the cavity 70 or72. The no-flow or low-flow characteristics of the partially cured epoxyprevents or minimizes epoxy flow into the cavities 70 or 72.

Proceeding to step 108, the ground plates 20 and 22 are each formed andattached to a spacer 16 or 18. In one embodiment, each ground plate 20and 22 is laminated to the respective spacer 16 or 18 using the no-flowor low-flow B-stage process previously described in connection with step106. The plates 20 and 22, in connection with the spacers 16 and 18 andthe support layer 12, form the propagation cavities 70 and 72.

Next, at step 110, the laminated layers are pressed together and heatedto cure the epoxy and form the basic structure of the suspendedtransmission line 10. For the low-flow B-stage process, the layers maybe pressed together at a pressure of 250 -300 psi and heated at atemperature of 350 degrees Fahrenheit for 90 minutes.

At step 112, the mode suppression connectors 68 are formed for thesuspended transmission line 10. In one embodiment, the mode suppressionconnectors 68 are each formed by drilling a via through the plates 20and 22, spacers 16 and 18, and support layer 12, and plating the viaswith copper. In this embodiment, the vias preferably have a diametergreater than 0.04 inches to allow copper and tin plating through theentirety of the vias. In this way, the suspended transmission line 10 isefficiently fabricated further reducing transmission line cost. Inaddition, conventional multi-layer printed circuit board fabricationtechniques may be used in fabrication of the suspended transmission line10 eliminating the need for the development and testing of newtechniques and equipment.

FIG. 4 illustrates a method for transmitting a signal in the suspendedtransmission line 10. The method begins at step 114 in which theconductor 114 is supported in the low-loss propagation structure 50. Aspreviously described, the low-loss propagation structure includes aircavities 70 and 72 formed above and below the conductor 14.

Proceeding to Step 115, a signal is transmitted along the conductor 14.At step 116, an electric field generated by the signal is substantiallycontained to the low-loss propagation structure 50. In addition, at step118, a substantially constant phase and amplitude for the signal ismaintained in the conductor 14. Accordingly, line losses are minimizedin the suspended transmission line 10.

FIG. 5 illustrates a suspended transmission line segment 120 having anembedded signal channeling device 122 in accordance with one embodimentof the present invention. In this embodiment, the suspended transmissionline segment 120 and embedded signal channeling device 122 form a powerdivider 124 with a radio frequency (RF) input port 126 and a pluralityof RF output ports 128 that allow connection into existing transmissionsystems with coaxial cable. Use of the power divider 124 substantiallyeliminates design, engineering, construction, and maintenance costsassociated with a separate mechanical housing for the signal channelingdevice 122. Further use of the suspended transmission line structure asthe transmission line substantially eliminates the need for the RFconnectors 126 and 128 and further reduces cost while improvingperformance by minimizing line losses associated with the RF connectors.

Referring to FIGS. 5 and 6, the transmission line segment 120 includes asupport layer 130, first and second spacers 132 and 134, and first andsecond plates 136 and 138. In one embodiment, the support, spacer, andplate layers 130, 132, 134, 136, and 138 are fabricated from materialsand configured as previously described in connection with correspondinglayers of the suspended transmission line 10. The various parts of thetransmission line segment 120 are laminated together and form a low-losspropagation structure 140 enclosing a conductor 142 supported by thesupport layer 130. The conductor 142 is an integrated broadsideconductor and extends from the RF inlet port 126 to the RF outlet ports128. Mode suppression connectors are formed as also described inconnection with the suspended transmission line 10.

For the illustrated embodiment, the signal channeling device 122 is afloating node three-way Wilkinson divider. In this embodiment, thesignal channeling device 122 includes a combined signal line 150, threediscrete signal lines 152, and an interference blocking system 154. Thediscrete signal lines 152 extend from a junction 155 with the combinedsignal line 150 and are each operable to transmit a portion of a signaltransmitted on the combined signal line 150. The combined and discretesignal lines 150 and 152 are formed by the conductor 142 and eachinclude a first conductive trace formed on a first side of the supportlayer 130, a second conductive trace formed on a second side of thesupport layer 130, and a plurality of connectors that each connect thefirst and second conductive traces at intermediate points along thelength of the line 150 or 152. The conductive traces and connectors areformed as previously described in connection with corresponding elementsof the center conductor 14 of the suspended transmission line 10.

The interference blocking system 154 absorbs downstream power backing upto the discrete signal lines 152. The interference blocking system 154includes, for each discrete signal line 152, a 50 Ohm or other suitablepower resistor 156 connecting the line 152 to a floating node 158. Thepower resistors 156 are preferably disposed on the outside of the firstplate 136 to provide good heat dissipation and accommodate the geometryof the suspended transmission line segment 120. In a particularembodiment, the power resistors 156 are each coupled to one of thediscrete signal lines 152 through a connector 160 and pad 180. Theconnectors 160 each extend from a power resistor 156 through the firstplate 136 and a supporting projection 162 of the first spacer 132 thatinterrupts the propagation structure 140. The interference blockingsystem 154 may comprise other suitable elements, be otherwise suitablyconfigured, or be otherwise suitably integrated with the suspendedtransmission line segment 120.

FIG. 6 illustrates details of the combined and discrete signal lines 150and 152 (FIG. 5) of the floating node three-way Wilkinson divider. TheWilkinson divider splits an input signal into three output signals.Second stage Wilkinson dividers are interconnected to further split eachoutput signal of an upstream divider into three additional signals.

Referring to FIG. 6, the combined signal line 150 extends from the inputport 126 to the junction 155 with the discrete signal lines 152. Thecombined signal line 150 includes a capacitive input section 170. Thecapacitive input section 170 tunes out the power resistor 156 (FIG. 5)parasitic capacitance at the front end of the power divider 124 (FIG.5).

The discrete signal lines 152 include first and second outside pathways172 and 174 and a center pathway 176. The pathways 172, 174, and 176 areconfigured and sized to provide a 90 degree or other suitable phaseshift of, and a 77 Ohm or other suitable impedance to, a signaltransmitted on the combined signal line 150. This is accomplished bymaintaining a constant impedance of approximately 77 Ohms . Theimpedance is not restricted to this value and may in some cases becloser to 86.6 Ohms for the pathways 172, 174 and 176. The outside andcenter pathways 172, 174, and 176 each extend from the junction 155 withthe combined signal line 150 to a separate RF output port 128 which hasa linewidth that produces an impedance of approximately 50 Ohms. Theimpedance of these lines maintains a low VSWR. The first and secondouter pathways 172 and 174 substantially mirror each other on oppositesides of the center pathway 176. The center pathway 176 includes aserpentine element 178 to maintain a length substantially equal to thatof the outside pathways 172 and 174. Equal length of the outside andcenter pathways 172, 174, and 176 produce a substantially constant phaseshift in each pathway. Accordingly, a signal transmitted on the combinedsignal line 150 is equally split between the pathways 172, 174, and 176.

The power resistors 156 (FIG. 5) are connected to each of the outsideand center pathways 172, 174, and 176 by the connectors 160 (FIG. 5) atnodes 180. As previously described, the power resistors 156 absorbdownstream power backing up to the pathways 172, 174, or 176.

In a particular embodiment, the power divider 124 is used to feed acellular antenna with multiple discrete antennas at a frequency of 800MHz. For a 100-watt conductor 142, the power resistors 156 are BeOresistors. BeO resistors may be used because of their excellent thermaldissipation when mounted on first plate 136 (FIG. 5). The combinedsignal stripline 150 has a line width of 44 mils and length of 57 milsfrom the input port 126 to the capacitive input section 170. Thecapacitive input section 170 has a width of 200 mils and a length of 150mils . From the stripline capacitive input section 170 to the junction155 with the discrete signal lines 152, the combined signal stripline150 has a line width of 76 mils and length of 100 mils. For the discretesignal suspended transmission lines 152, the outside and center pathways172, 174, and 176 each have a line width of 30 mils and a length of aquarter wavelength from the junction 155 with the combined signal line150 to the resistor connect nodes 180. The propagation structure 140(FIG. 5) comprises a first air cavity above the conductor 142 (FIG. 5)and a second air cavity below the conductor 142. Each air cavity has awidth of about 3 to 5 times line width of the outside and centerpathways 172, 174, and 176 mils and a height of 86.5 mils. From theresistor connect nodes 180, the pathways 172, 174, and 176 each extendto a separate one of the RF output ports 128. At the output ports 128,each divided signal has a frequency of 800 MHz. The power divider 124(FIG. 5) may be otherwise suitably configured or formed in a segment ofa suspended transmission line or in an extended run of a suspendedtransmission line.

FIG. 7 illustrates further details of the combined and discrete signallines 150 and 152 (FIG. 5) of the floating node 3-way Wilkinson dividerin accordance with one embodiment of the present invention. In thisembodiment, the input and output are 50 Ohms suspended transmission linefrom other circuitry or the input port 126 (FIG. 5) and output port 128(FIG. 5), respectively. The input and output structures are in striplinefor mechanical support. These structures are 50 Ohms, as short aspossible. In addition, periodic mechanical support is provided in thesuspended transmission lines. The periodic mechanical support comprisesfirst and second spacers 132 and 134 (FIG. 5). The arms are quarterwavelength and 77 Ohms due to minimum widths.

The support projection 162 (FIG. 5), in the areas for the resistor 156(FIG. 5), connector 160 (FIG. 5) and the pad 180, may be solid filledG-10 board. Stripline, solid filled G-10 board, also forms a supportingprojection for the areas of the input port 126 and the output port 128.The cavity area, which also demarcates the suspended transmission line,is illustrated as a dotted line. Additional periodic smaller supportsare shown in the cavity area to keep the line suspended as necessary.

The characteristic impedance of the different lines is labeled. Thecharacteristic impedance describes the electrical characteristics of thelines and determines the geometric proportion of the line elements:i.e., line width in proportion to cavity height. Therefore it can bescaled. Similarly, the line length is labeled in terms of number ofwavelengths. The discrete signal lines 152, each have a characteristicimpedance of 77 Ohms and a length of a quarter-wavelength for any givenfrequency. While, a characteristic impedance of 86.6 Ohms may bepreferred, it makes the lines too narrow for power handling. Thecopositive input section 170 has a characteristic impedance and lengthas required for compensating out parasitics and nonidealities (such as a77 Ohm line instead of an 86.6 Ohm line). All the other lines on theinputs and outputs are 50 Ohms and their lengths are as required.

FIG. 8 illustrates a method for dividing a signal in the power divider124. The method begins at step 200 to support the conductor 142 in thelow-loss propagation structure 140 of the suspended transmission linesegment 120. Next, at step 202, a signal is received at the input port126 and transmitted along the combined signal line 150 of the conductor142.

Proceeding to step 204, the signal is divided onto the discrete signallines 152 of the conductor 142. As previously described, the discretesignal lines 152 include outside pathways 172, 174, and a center pathway176. The pathways 172, 174, and 176 are each of substantially equallength and provide a substantially constant phase shift of thetransmitted signal. At step 206, the divided signals are eachtransmitted along one of the pathways 172, 174, or 176 to acorresponding RF output port 128. From the RF output ports 128, thedivided signals are each fed to an element of multiple antenna systems.

During operation, at step 208, an electromagnetic field generated by thetransmitted signal is substantially contained by, or to, the low-losspropagation structure 140. In addition, in each of the combined anddiscrete signal lines 150 and 152, a substantially constant phase andamplitude is maintained for the signal on the line 150 or 152.Accordingly, line losses are minimized in the power divider 124. Step210 leads to the end of the process by which the transmitted signal isdivided for feed to multiple elements while minimizing line losses.

Although the present invention has been described with severalembodiments, various changes and modifications may be suggested to oneskilled in the art. It is intended that the present invention encompasssuch changes and modifications as fall within the scope of the appendedclaims.

What is claimed is:
 1. An embedded signal channeling device for a transmission line, comprising: a dielectric support layer having a first side and a second side; a combined signal line supported by the support layer and having a first part supported by the first side of the support layer and a second part supported by the second side of the support layer; a plurality of discrete signal lines supported by the support layer, each discrete signal line having a first part supported by the first side of the support layer and a second part supported by the second side of the support layer, the plurality of discrete signal lines extending from the combined signal line and each transmitting a portion of a signal transmitted on the combined signal line; a first ground plane positioned from the first side of the dielectric support layer in a spaced relationship, the spacing between the first ground plane and the first side selected to form a propagation structure to substantially contain the electromagnetic field generated by a transmitted signal on the combined signal line and the plurality of discrete signal lines in the propagation structure; a second ground plane positioned from the second side of the dielectric support layer in a spaced relationship, the spacing between the second ground plane and the second side selected to form a propagation structure to substantially contain the electromagnetic field generated by a signal transmitted on the combined signal line and the plurality of discrete signal lines in the propagation structure; and a plurality of broadside connectors spaced at substantially equal distances along the combined signal line and each of the plurality of discrete signal lines to maintain a substantially constant phase and amplitude for the signal transmitted on each of the lines, the plurality of broadside connectors connecting the first and second parts along the length of each of the signal lines.
 2. The embedded signal channeling device of claim 1, wherein each of the plurality of discrete signal lines comprise a length imparting a substantially equal phase to the signal.
 3. The embedded signal channeling device of claim 2, wherein each of the plurality of discrete signal lines have substantially the same length.
 4. The embedded signal channeling device of claim 3, wherein the discrete signal lines further comprise: a first outside line; a second outside line; and a center line between the first and second lines and including a serpentine element to maintain substantially the same length as the first and second outside lines.
 5. The embedded signal channeling device of claim 1, further comprising for each of the plurality of discrete signal lines, a power resistor coupled to the discrete signal line to absorb downstream interference on the discrete signal line.
 6. The embedded signal channeling device of claim 5, further comprising the power resistors supported on an outside of the first ground plane and each coupled to a corresponding discrete signal line by a connector extending through the first ground plane to the corresponding discrete signal line.
 7. The embedded signal channeling device of claim 1, wherein the support layer comprises a lossy material.
 8. The embedded signal channeling device of claim 7, wherein the lossy support layer comprises an epoxy glass material.
 9. The embedded signal channeling device of claim 1, wherein the combined signal line and the discrete signal lines comprises a first metalization supported on a first side of the support layer and a second metalization supported on a second side of the support layer.
 10. The embedded signal channeling device of claim 9, wherein the first and second metalizations substantially mirror each other.
 11. The embedded signal channeling device of claim 9, further comprising a plurality of connectors extending between the first and second metalizations at intermediate points along the length of the combined and discrete signal lines.
 12. The embedded signal channeling device of claim 11, wherein the connectors are substantially evenly spaced along the length of the combined and discrete signal lines.
 13. The embedded signal channeling device of claim 1, wherein the propagation structure comprises a first air cavity between the combined and discrete signal lines and the first ground plane and a second air cavity between the combined and discrete signal lines and the second ground plane.
 14. The embedded signal channeling device of claim 13, further comprising: a first spacer between the support layer and the first ground plane to form the first air cavity between the combined and discrete signal lines and the first ground plane; and a second spacer between the support layer and the second ground plane to form the second air cavity between the combined and discrete signal lines and the second ground plane.
 15. The embedded signal channeling device of claim 14, further comprising a plurality of mode suppression connectors connecting the first and second ground planes at spaced intervals.
 16. An embedded signal channeling device for a transmission line, comprising: a support layer; a conductor supported on the support layer between first and second plates, each having a ground plane, the conductor including a combined signal line and a plurality of discrete signal lines extending from the combined signal line and operable to each transmit a portion of a signal transmitted on the combined signal line; a propagation structure between the first and second plates to substantially contain and electromagnetic field generated by the signal; and a power resistor for each of the plurality of discrete signal lines coupled to the discrete signal line to absorb downstream interference on the discrete signal line, each power resistor supported on the first plate and each coupled to a corresponding discrete signal line by a connector extending through the first plate to the corresponding discrete signal line.
 17. The embedded signal channeling device of claim 16, wherein each of the plurality of discrete signal lines comprise a length imparting a substantially equal phase to the signal.
 18. The embedded signal channeling device of claim 17, wherein the discrete signal lines further comprises: first outside line; a second outside lines; and a center line between the first and second lines and including a serpentine element to maintain substantially the same length as the first and second outside lines.
 19. The embedded signal channeling device of claim 16, further comprising for each of the plurality of discrete signal lines, a power resistor coupled to the discrete signal line to absorb downstream interference on the discrete signal line.
 20. The embedded signal channeling device of claim 19, further comprising the power resistors supported on an outside of the first plate and each coupled to a corresponding discrete signal line by a connector extending through the first plate to the corresponding discrete signal line.
 21. The embedded signal channeling device of claim 16, wherein the combined signal line and the discrete signal lines comprises a first metalization supported on a first side of the support layer and a second metalization supported on a second side of the support layer.
 22. The embedded signal channeling device of claim 21, further comprising a plurality of connectors extending between the first and second metalizations at intermediate points along the length of the combined and discrete signal lines.
 23. The embedded signal channeling device of claim 16, wherein the propagation structure comprises a first air cavity between the conductor and the first plate and a second air cavity between the conductor and the second plate.
 24. The embedded signal channeling device of claim 23, further comprising: a first spacer between the support layer and the first plate to form the first air cavity between the conductor and the first plate; and a second spacer between the support layer and the second plate to form the second air cavity between the conductor and the second plate.
 25. The embedded signal channeling device of claim 16, further comprising a plurality of mode suppression connectors connecting the first and second plates at spaced intervals. 